Power Transfer Devices, Methods, and Systems with Crowbar Switch Shunting Energy-Transfer Reactance

ABSTRACT

The present application discloses methods, circuits and systems for power conversion, using a universal multiport architecture. When a transient appears on the power input (which can be, for example, polyphase AC), the input and output switches are opened, and a crowbar switch shunts the inductance which is used for energy transfer. This prevents this inductance from creating an overvoltage when it is disconnected from outside lines.

CROSS-REFERENCE

Priority is claimed both from U.S. application 61/221,345 filed Jun. 29,2009 and from U.S. application 61/234,373 filed Aug. 17, 2009. Priorityis also claimed from U.S. application Ser. No. 12/479,207, filed Jun. 5,2009 and now published as US2010/0067272, and therethrough from U.S.application Ser. No. 11/759,006, filed Jun. 6, 2007 and now issued as7,599,196, and therethrough from U.S. application 60/811,191 filed Jun.6, 2006. Priority is also claimed from U.S. application Ser. No.11/758,970 filed Jun. 6, 2007, and also therethrough from U.S.application 60/811,191 filed Jun. 6, 2006. All of these applications arehereby incorporated by reference.

BACKGROUND

The present application relates to power converters, and moreparticularly to non-resonant power converters which use an inductivereactance for energy transfer.

Note that the points discussed below may reflect the hindsight gainedfrom the disclosed inventions, and are not necessarily admitted to beprior art.

Power conversion is one of the most important applications of powersemiconductors, and plays an important role in many systems. Powerconversion can be used to shift the voltage of a power supply to matchthe operating requirements of a particular load, or to permit use of avariable-voltage or variable current supply, or to compensate for thevariation expected in an unreliable power source, or to permit a unit tobe usable with a variety of power inputs, or to compensate for shift in“power factor” when an AC supply is connected to a reactive load. Inmany cases there are different terms for particular kinds of powerconversion, e.g. a DC-to-AC converter is often referred to as aninverter, and some types of AC-to-AC converter are referred to ascycloconverters. Many kinds of motor drive can be thought of as a kindof power conversion: for example, a variable-frequency drive can beregarded as a species of power converter in which the frequency of an ACoutput is adjustable. In the present application the term “powerconversion” will be used to refer generically to all of these types.

The present inventor has previously filed on a new class of powerconverter device operation and device, which provides a nearly universalpower conversion architecture. In one version of this architecture, eachinput line is connected to the middle of one phase leg having twobidirectional switches, and the switches are operated so as to drive theterminals of a link reactance from one input or the other. Acorresponding output switch array is used to transfer energy from thelink reactance into two or more output terminals as desired, toconstruct the output waveform desired. Preferably the link reactanceincludes an inductor which is shunted by a capacitor. This provides anearly universal hardware architecture, which is operated to implement adesired power-conversion function. This architecture is suitable forDC-AC, AC-AC, and AC-DC conversion. However, the present inventor nowprovides additional improvements, which are applicable to these as wellas other topologies.

Many DC-DC, DC-AC, and AC-AC Buck-Boost converters are shown in thepatent and academic literature. The classic Buck-Boost converteroperates the inductor with continuous current, and the inductor may havean input and output winding to form a transfer for isolation and/orvoltage/current translation, in which case it is referred to as aFlyback Converter. There are many examples of this basic converter, allof which are necessarily hard switched and therefore do not have thesoft-switched attribute, which leads to reduced converter efficiency andhigher costs. An example of a hard switched three phase to three phaseBuck-Boost converter is shown in FIG. 4, from K. Ngo, “Topology andAnalysis in PWM Inversion, Rectification, and Cycloconversion,”Dissertation, California Institute of Technology (1984).

One proposed DC-AC Buck-Boost converter (in U.S. Pat. No. 5,903,448)incorporates a bi-directional conduction/blocking switch in its outputsection to accommodate four quadrant operation, with AC output andbi-directional power transfer. The input, however, cannot be AC, and ituses hard switching.

SUMMARY

The present application discloses new approaches to transient voltagesin a link-type power conversion circuit. In one class of preferredembodiments, semiconductor switches not only provide connection of eachinput line to either end of the link reactance, but anothersemiconductor switch shunts the link reactance to prevent overvoltageswhen the input/output ports are shut off.

The disclosed innovations, in various embodiments, provide one or moreof at least the following advantages. However, not all of theseadvantages result from every one of the innovations disclosed, and thislist of advantages does not limit the various claimed inventions.

The following advantages are provided by the inventive embodiment ofFIG. 1, and by many other disclosed and equivalent embodiments:

-   -   Low harmonic, unity power factor current draw from the utility,        regardless of output voltage.    -   Current is drawn from each phase in high frequency pulses,        similar to a current source converter, with input capacitors and        optionally, line inductors, converting the pulsed current flow        to sinusoidal current flow.    -   Ability to step up or step down voltage from input to output,        allows full output voltage even in the presence of input voltage        sags, as commonly occurs in industrial power systems.    -   Sinusoidal output voltage with small voltage ripple allows        standard induction motors, as well as low reactance synchronous        motors, to be used.    -   Output capacitors filter the pulsed current.    -   Ripple frequency is always high so as to avoid any resonance        problems with input and/or output filters or reactances.    -   Ability to supply 200% or higher of nominal output current at        low output voltages, indefinitely, as may be advantageous for        starting large inertial loads.    -   With near zero output voltages, the converter is operated at        about half or maximum frequency, with the inductor first fully        charged by the input, then discharging at that full level into        the output for twice the full voltage discharge period, then        discharging to zero current back into the input, repeating that        cycle but with reverse current.    -   Peak currents remain the same, but output current is doubled.    -   Input-Output isolation, resulting in zero common mode voltages        on the output.    -   Since there is never a moment when the input and output lines        are connected together, as happens continuously in voltage and        current source drives, as well as matrix converters, the average        output voltage remains at ground potential. This eliminates the        need for isolation transformers.    -   Slow reverse recovery devices are usable. Rate of change of        current during commutation is relatively slow, and applied        reverse voltage after reverse recovery is also low, so the        switches used may have rectifier diode like recovery        characteristics. This is particularly advantageous with reverse        blocking IGBTs and GTOs, which are inherently slow to reverse        recover.    -   Slower forward turn-off devices are usable.    -   Turn-off dv/dt is relatively low due to the capacitance in        parallel with the inductor.    -   Compact, lightweight, and efficient.    -   Conventional voltage source drives require multiple heavy and        bulky power inductors, one on each of the input and output        lines. For a given output quality, this is not true of the        innovative embodiments disclosed here.    -   Conventional current source drives require a very large and        heavy DC inductor in order to generate full output voltage. For        a given output quality, this is not true of the innovative        embodiments disclosed here.    -   Weight can be reduced by a factor of 10, as compared to a        suitably filtered, commercially available voltage source drive        for 40 hp is over 300 pounds, while the drive of this invention        will weigh less than 30 lbs for 40 hp.    -   Lack of large input/output filter inductors significantly        improves the efficiency of this invention over conventional        drives.    -   No transformers are needed since input current harmonics are low        and there is no common mode output voltage.    -   The parts count is relatively small.    -   By using bi-directional switches, only 12 power switches are        needed for a polyphase drive.    -   High bandwidth. Since the current amplitude is determined twice        each cycle of the inductor, the current control bandwidth of        this invention is inherently very high, making the invention        suitable for high bandwidth servo applications and even high        power audio amplifiers.

BRIEF DESCRIPTION OF THE DRAWINGS

The disclosed inventions will be described with reference to theaccompanying drawings, which show important sample embodiments and whichare incorporated in the specification hereof by reference, wherein:

FIG. 1 shows a sample embodiment in a Full-Bridge Buck-Boost Converter.

FIGS. 2 a-2 d show four alternative versions of the basic Bi-directionalConducting and Blocking Switch (BCBS) used in the circuit of FIG. 1.

FIG. 3 shows a conventional “Standard Drive”.

FIG. 4 shows a conventional hard-switched three phase to three phase ACbuck-boost converter.

FIG. 5 shows a conventional soft-switched “partial resonant” three phaseto three phase AC buck-boost converter, and FIG. 6 shows the inductorcurrent and voltage waveforms for the converter of FIG. 5.

FIGS. 7 and 8 show other conventional converters.

FIG. 9 shows the input line voltages, and FIG. 10 shows the output linevoltages, for the current switching example of FIGS. 11, 12 a-12 j, and13.

FIG. 11 summarizes the line and inductor current waveforms for severalinductor cycles.

FIGS. 12 a-12 j show voltage and current waveforms on the inductorduring a typical cycle.

FIG. 13 shows voltage and current waveforms corresponding to the fullpower condition of FIG. 12, and FIG. 14 shows inductor voltage andcurrent for an output voltage of about half the full output voltage.

FIG. 15 shows another embodiment, which includes Controls and I/OFiltering.

FIG. 16 illustrates current and timing relationships in yet otherembodiments, and FIG. 17 shows these relationships when the outputvoltage is half of the input voltage.

FIG. 18 is a spreadsheet which calculates the average output current fora given set of conditions, as the current discharge time is varied, andFIG. 19 shows the results of this calculation under various conditions.

FIG. 20 is a version of FIGS. 16 and 17 which shows inductor current andtiming for a regeneration condition where the output voltage is ½ of theinput.

FIG. 21 shows yet another embodiment, with DC or Single Phase portals.

FIG. 22 shows yet another embodiment, with a Transformer.

FIG. 23 shows yet another embodiment, in a four portal configuration.

FIG. 24 shows yet another embodiment, in a three portal applicationmixing three phase AC portals and a DC portal, as may be used toadvantage in a Hybrid Electric Vehicle application.

FIGS. 25 and 26 show two more classes of implementations usinghalf-bridge topologies.

FIG. 27 shows yet another embodiment, which provides a single phase tothree phase synchronous motor drive.

FIG. 28 shows yet another embodiment, with dual power modules.

FIG. 29 shows yet another embodiment, which provides a three phase PowerLine Conditioner.

DETAILED DESCRIPTION OF SAMPLE EMBODIMENTS

The numerous innovative teachings of the present application will bedescribed with particular reference to presently preferred embodiments(by way of example, and not of limitation). The present applicationdescribes several inventions, and none of the statements below should betaken as limiting the claims generally.

The present application discloses power converters which are generallyof the Buck-Boost family, but which use capacitance, either parasiticalone or with added discrete device(s), in parallel with the Buck-Boostinductor to achieve low turn-off switching stresses (i.e. “softswitching”) on the semiconductor switches, allowing relatively slow andinexpensive switches to be used. In alternative disclosed embodiments,as discussed below, operation without such added capacitance ispossible, at the expense of higher forward turn-off switching losses.

In FIG. 1, and various other disclosed embodiments, even if little or noparallel capacitance is used, switch turn on always occurs as the switchtransitions from reverse to forward bias, allowing for low turn-onlosses. Reverse recovery of the switches is accomplished with low ratesof current decrease, and with low reverse recovery voltage, leading tonear zero loss reverse recovery switching.

The embodiments described below are improvements on the “Full Cycle”mode of the parent applications, which results in two power transfersper inductor cycle. Buck-Boost converters, including those of the Ngoand Kim references cited above, have a DC bias in the inductor current,and only one power transfer per inductor cycle.

The disclosed inventions can also be used for DC-AC, AC-DC, AC-AC, orDC-DC conversion, with no limitation on the relative magnitudes of thevoltages involved as long as the voltage rating of the switches is notexceeded. However, if the implementation is such that one portal isalways a higher voltage than the other portal, then the switchesconnected to said higher portal need only be able to block voltage inone direction.

Full electrical isolation and/or greater voltage and current conversionmay be achieved by using an inductor/transformer instead of the simpleinductor. Note that the inductor/transformer will typically not havecurrent in both sides at the same time, so its operation is more like asplit inductor (as in a flyback converter) than like a simpletransformer (as in a push-pull converter. Another significant differentbetween buck-boost and push-pull is that the push-pull output voltage isfixed as a multiple or fraction of the input voltage, as given by theturns ratio, while the buck-boost has no such limitation. Push-pulltopologies are described athttp://en.wikipedia.org/wiki/Push-Pull_Converter, which (in its state asof the filling date) is hereby incorporated by reference. A push-pull isquite unlike a buck-boost or flyback converter, in that the transformeris not operated as an energy-transfer inductor. In a buck-boost orflyback, input current pumps energy into a magnetic field, which is thendrained to drive output current; thus the input and output currents flowat different times.

Inductor/transformer leakage inductance is typically a significantconcern of buck-boost designs. This is typically dealt with byminimizing the leakage, and sometimes by adding circuit elements to dealwith it. By contrast, many of the embodiments described below cantolerate large parasitic capacitance, and thus inductors or transformerswith very close windings can be specified, to minimize the leakageinductance. The standard hard switched buck-boost cannot tolerateparasitic capacitance, which makes it very difficult to minimize theleakage inductance for those configurations.

The innovative converter circuits, in various elements are constructedof semiconductor switches, an inductor, advantageously a capacitor inparallel with the inductor, and input and output filter capacitances. Acontrol means, controlling the input switches, first connects theinductor, initially at zero current, to the input voltage, which may beDC or the highest line-to-line voltage AC pair in a three phase input,except at startup, in which case a near zero-voltage line pair is used.The control then turns off those switches when the current reaches apoint determined by the control to result in the desired rate of powertransfer. The current then circulates between the inductor andcapacitor, which results in a relatively low rate of voltage increase,such that the switches are substantially off before the voltage acrossthem has risen significantly, resulting in low turn-off losses.

With DC or single phase AC input, no further current is drawn from theinput. With three phase AC input, the control will again connect theinductor to the input lines, but this time to the line-to-line pairwhich has a lower voltage then the first pair. Turn on is accomplishedas the relevant switches transition from reverse to forward bias. Afterdrawings the appropriate amount of charge (which may be zero if thecontrol determines that no current is to be drawn from the pair, as forexample that the pair is at zero volts and input unity power factor isdesired), the relevant switches are again turned off. Under mostconditions, the voltage on the inductor will then reverse (withrelatively low rates of voltage change due to the parallel capacitance).With three phase AC output, the control will turn on switches to allowcurrent to flow from the inductor to the lowest voltage pair of lineswhich need current, after the relevant switches become forward biased,with the control turning off the switches after the appropriate amountof charge has been transferred. The inductor voltage then ramps up tothe highest output line-to-line pair for three phase AC, or to theoutput voltage for single phase AC or DC. Again, switches are turned onto transfer energy (charge) to the output, transitioning from reverse toforward bias as the voltage ramps up. If the output voltage is largerthen the highest input voltage, the current is allowed to drop to zero,which turns off the switch with a low rate of current reduction, whichallows for the use of relatively slow reverse recovery characteristics.If the output voltage is less than the highest input voltage, theswitches are turned off before current stops, so that the inductorvoltage ramps up to the input voltage, such that zero-voltage turn on ismaintained. Alternatively, the switches may be turned off before thepoint cited in the previous sentence, so as to limit the amount ofcurrent into the output. In this case, the excess energy due to currentin the inductor is directed back into the input by turning on switchesto direct current flow from the inductor into either the highest voltagepair in three phase, or the single phase AC or DC input.

In a three phase AC converter, the relative charge per cycle allocatedto each input and output line pair is controlled to match the relativecurrent levels on each line (phase). After the above scenario, when zerocurrent is reached the inductor is reconnected to the input, but with apolarity reversed from the first connection, using switches that arecomplimentary to the switches used in the first half of the cycle. Thisconnection can occur immediately after zero current (or shortly afterzero current if the input voltage is less than the output voltage, toallow the capacitor voltage time to ramp back down), giving fullutilization of the power transfer capability of the inductor. Noresonant reversal is required as in the time period M4 of the Kimconverter shown in FIGS. 5 and 6.

The disclosed embodiments are inherently capable of regeneration at anycondition of output voltage, power factor, or frequency, so in motordrive or wind power applications, the motor may act as a generator,returning power to the utility lines.

In an AC motor drive implementation, input and output filtering may beas little as line-to-neutral connected capacitors. Since switches lossesare very low due to soft switching, the Buck-Boost inductor can beoperated at a high inductor frequency (typically 5 to 20 kHz for lowvoltage drives), allowing for a single, relatively small, and low loss,magnetic device. The current pulse frequency is twice the inductorfrequency. This high frequency also allows the input and output filtercapacitors to be relatively small with low, high frequency ripplevoltage, which in turns allows for small, low loss line reactors.

Input voltage “sags”, as are common when other motors are connectedacross the line, are accommodated by temporarily drawing more currentfrom the input to maintain a constant power draw and output voltage,utilizing the boost capability of this invention, avoiding expensiveshutdowns or even loss of toque to the application.

The full filter between the converter and an attached voltage source(utility) or sink (motor, another utility, or load) includes the linecapacitance (line-to-line or line-to-neutral, as in Y or Delta), and aseries line inductance (or “line reactor”). When driving a motor, theline reactance is just the inductance of the motor. This provides apower filter, AND does important conditioning for the converter.

The preferred converter benefits from having very low impedance voltagesources and sinks at the inputs and outputs. (This is a significantdifference from the converter of FIG. 7, which has inductive linereactance at the I/O, not capacitive.) The link inductor current must beable to be very rapidly switched between the link capacitor and the I/Ocapacitors, and line reactance would prevent that from incurring, and infact would likely destroy the switches. The physical construction of theconverter should preferably be carefully done to minimize all suchinductance which may impair link reactance switching.

The line capacitance itself does not have to have any particular value,but for proper operation the change in voltage on the line capacitancewhile charging or discharging the link inductance should only be a smallfraction of the initial voltage, e.g. less than 10%. There are otherrestraints as well. For a 20 hp, 460 VAC prototype, 80 μF ofline-to-neutral capacitance results in only a 1 to 2% ripple voltage.(This large capacitance was chosen in order to get the ripple currentwithin the capacitor's current rating.) Capacitors could be made withlower μF for the same current rating, resulting in smaller, cheapercapacitors, and higher voltage ripple, but this is all that is availableright now.

Another important consideration is the resonant frequency formed by theL-C of the line reactance and the line capacitance (the I/O powerfilter). This frequency must be lower than the link power cyclefrequency in order to not have that filter resonant with the voltageripple on the line capacitance. For the 20 hp 460 VAC prototype example,the link frequency was 10 kHz, so the link power cycle frequency is 20kHz (2 power cycles per link voltage cycle). Since the resonantfrequency of the L-C I/O is lower than 2 kHz, that works well.

So, to summarize, the capacitance needs to be large enough to reasonablystabilize the I/O voltage to allow the link inductor charge/discharge tooccur properly, and the L-C resonant frequency needs to be less thantwice the link voltage frequency, and generally at least 4 to 10 timeslower.

It should also be noted that too much capacitance on a line filter canlead to excess reactive power on the utility connection.

Referring initially to FIG. 1, illustrated is a schematic of a threephase converter 100 as taught by the present application. The converter100 is connected to first and second power portals 122 and 123. Each ofthese portals can source or sink power, and each has a port for eachphase of the portal. Converter 100 serves to transfer electric powerbetween the portals, and can accommodate a wide range of voltages,current levels, power factors, and frequencies. The first portal can be,for example, a 460 VAC 60 Hz three phase utility connection, while thesecond portal can be e.g. a three phase induction motor which is to beoperated at variable frequency and voltage so as to achieve variablespeed operation of said motor. This configuration can also accommodateadditional portals on the same inductor, as may be desired toaccommodate power transfer to and from other power sources and/or sinks.Some examples of these alternatives are shown in FIGS. 23 and 24, butmany others are possible.

The converter 100 includes a first set of electronic switches (S_(1u),S_(2u), S_(3u), S_(4u), S_(5u), and S_(6u)) which are connected betweena first port 113 of a link inductor 120 and each respective phase (124through 129) of the input portal. A second set of electronic switches(S_(1l), S_(2l), S_(3l), S_(4l), S_(5l), and S_(6l)) are similarlyconnected between a second port 114 of link inductor 120 and each phaseof the output portal. A link capacitor 121 is connected in parallel withthe link inductor, forming the link reactance. In this example, each ofthe switches in the switching array is capable of conducting current andblocking current in both directions, and may be composed of thebi-directional IGBT 201 of FIG. 2 b, as shown in U.S. Pat. No.5,977,569. Many other such bi-directional switch combinations arepossible, such as anti-parallel reverse blocking IGBTs 200 of FIG. 2 a,or the device configurations of FIGS. 2 c and 2 d.

Most of these switch combinations contain two independently controlledgates, as shown with all the switches for FIGS. 2 a-2 d, with each gatecontrolling current flow in one direction. In the following description,it is assumed that two gate switches are used in each switch, and thatthe only gate enable in a switch is the gate which controls current inthe direction which is desired in the subsequent operation of theswitch. Thus, when each switch mentioned below is said to be enabled,said enabling occurs before conduction occurs, since that portion of theswitch is reverse biased at the instant of being enabled, and does notconduct until it becomes forward biased as a result of the changingvoltage on the parallel pair of inductor and capacitor. Any switchembodiment which has only one gate, such as a one-way switch embeddedwithin a full wage bridge rectifier, must be enabled only when thevoltage across it is very small, which required precise and accuratetiming that may be difficult to achieve in practice.

Note that a link switch 199, with its series resistance 198, isconnected to shunt the energy transfer reactance. When a transientovervoltage appears on an external line which is currently connected tothe energy transfer reactance, it may be necessary to turn off theswitch connecting that voltage to the energy transfer reactance, toavoid overvoltages on the reactance. However, if the reactance isisolated from the external lines while it contains significant energy,it is possible for this stored energy itself to create an overvoltage onthe reactance.

FIG. 1A shows waveforms which help to illustrate the need for the linkswitch 199. If a voltage transient occurs on an external line (as shownin the waveforms of the first line), a voltage-sensing input willtrigger the control logic to turn on the link switch 199, and to turnoff all of the switches in the switch array where the transient occurs.(Preferably all switches in all the switch arrays are turned off at thistime.)

The second line of FIG. 1A shows that the voltage on the reactancedeclines sharply when the link switch 199 is turned on. The third lineof FIG. 1A shows that the current circulating through the reactance andthe link switch declines more slowly. (However, overvoltage is theprimary risk to be avoided here.)

The rate of change of the current on the link reactance will be affectedby the back voltage on the series resistance 198. The selection of thisresistance will be discussed below. Note too that the series resistancewill include the parasitic resistance of the switching device which isused for the link switch; if the device used includes enough seriesresistance, it may not be necessary to include a separate resistor.

The fourth line of FIG. 1A shows the effect of the reduced voltage onthe reactance: the voltage seen on at least some of the switches isreduced when voltage on the link reactance is reduced.

This provides an advantage which can be used in several ways: for agiven switch rating, a higher voltage maximum on the input can beaccepted. Alternatively, for a given line voltage specification,transient overvoltage tolerance can be improved, and/or looser devicespecifications can be used.

FIG. 1B shows what might occur, under the same conditions as FIG. 1A, ifthe link switch were not present. In this case the rise of the voltageon the link reactance is not interrupted, and hence the voltage on theI/O devices can rise to a level where they break down.

FIG. 1C shows the simple logic used, in this sample embodiment, toactivate the link switch. The control circuitry consists of programmingto monitor the input and output line-line and line-ground voltages, andif they exceed some limit, all I/O switches are turned off, while thelink switch is turned on. Even with the maximum link current flowing,this limits the link voltage to some sufficiently low voltage, e.g. 600volts or less (5 ohms with 120 amps max link current).

With all I/O switches off, the total line-line voltage may go as high as2× the switch voltage rating minus the link voltage, or as anexample—2×1200 volts−600 volts=1800 volts.

This is 600 volts higher than what a standard converter can withstandline-line with 1200 volt switches. This would be on 480 VAC lines, whichhave normal peak line-line volts of about 700 volts, but transients canforce that to over 1400 volts. Metal Oxide Varistors (MOVs) can limitthe voltage to less than 1600 volts, so the converter of FIG. 1 is safeagainst transients. However, conventional converters (or inverters) arenot, since the MOVs cannot limit the voltage to less than 1200 volts.

Strictly speaking, the function of the link switch and resistor is tolimit the link voltage to a relatively low level while the linetransient is occurring, but the link switch and resistor as aconsequence of being turned on also dissipate at least some link energy,and typically all of it. Certainly if SCRs were used, all link energywould be dissipated.

The link switch just has to be able to handle the peak link current fora few hundred microseconds, and does not have to turn on that current.It could be composed of SCRs in anti-parallel, which would allowrelatively small SCRs to be used. SCRs can be used since the converterdoes not have to be turned back on until the link switch current goes tozero. The switches don't have to be heat sunk, so mounting them is easy.

The converter 100 also has input and output capacitor filters 130 and131, respectively, which smooth the current pulses produced by switchingcurrent into and out of inductor 120. Optionally, a line reactor 132 maybe added to the input to isolate the voltage ripple on input capacitorfilter 131 from the utility and other equipment that may be attached tothe utility lines. Similarly, another line reactor, not shown, may beused on the output if required by the application.

For example, in one sample implementation, a 20 hp (15 kW) VariableFrequency Drive (VFD) was configured with a peak link current of 110 A.The link switch resistor, in this example, was rated for 5 ohms, 1000V,and 5 watts. The required link switch voltage capability, in thisexample, was specified at 1200 V.

For illustration purposes, assume that power is to be transferred in afull cycle of the inductor/capacitor from the first to the secondportal, as is illustrated in FIG. 13. Also assume that, at the instantthe power cycle begins as shown in FIG. 9, phases A_(i) and B_(i) havethe highest line to line voltage of the first (input) portal, linkinductor 120 has no current, and link capacitor 121 is charged to thesame voltage as exists between phase A_(i) and B_(i). The controllerFPGA 1500, shown in FIG. 15, now turns on switches S_(1u) and S_(2l),whereupon current begins to flow from phases A_(i) and B_(i) into linkinductor 120, shown as Mode 1 of FIG. 12 a. FIG. 13 shows the inductorcurrent and voltage during the power cycle of FIG. 12 a-12 j, with theConduction Mode sequence 1300 corresponding to the Conduction Modes ofFIGS. 12 a-12 j. The voltage on the link reactance remains almostconstant during each mode interval, varying only by the small amount thephase voltage changes during that interval. After an appropriate currentlevel has been reached, as determined by controller 1500 to achieve thedesired level of power transfer and current distribution among the inputphases, switch S_(2l) is turned off.

Current now circulates, as shown in FIG. 12 b, between link inductor 120and link capacitor 121, which is included in the circuit to slow therate of voltage change, which in turn greatly reduces the energydissipated in each switch as it turns off. In very high frequencyembodiments of this invention, the capacitor 121 may consist solely ofthe parasitic capacitance of the inductor and/or other circuit elements.

To continue with the cycle, as shown as Mode 2 of FIG. 12 c and FIG. 13,switch S_(3l) is next enabled, along with the previously enabled switchS_(1u). As soon as the link reactance voltage drops to just less thanthe voltage across phases A_(i) and C_(i), which are assumed for thisexample to be at a lower line-to-line voltage than phases A_(i) andB_(i), as shown in FIG. 9, switches S_(1u) and S_(3l) become forwardbiased and start to further increase the current flow into the linkinductor, and the current into link capacitor temporarily stops. The two“on” switches, S_(1u) and S_(3l), are turned off when the desired peaklink inductor current is reached, said peak link inductor currentdetermining the maximum energy per cycle that may be transferred to theoutput. The link inductor and link capacitor then again exchangecurrent, as shown in FIG. 12 b, with the result that the voltage on thelink reactance changes sign, as shown in graph 1301, between modes 2 and3 of FIG. 13.

Now as shown in FIG. 12 d, output switches S_(5u) and S_(6l) areenabled, and start conducting inductor current into the motor phasesA_(o) and B_(o), which are assumed in this example to have the lowestline-to-line voltages at the present instance on the motor, as shown inFIG. 10. After a portion of the inductor's energy has been transferredto the load, as determined by the controller, switch S_(5u) is turnedoff, and S_(4u) is enabled, causing current to flow again into the linkcapacitor, which increases the link inductor voltage until it becomesslightly greater than the line-to-line voltage of phases A_(o) andC_(o), which are assumed in this example to have the highestline-to-line voltages on the motor, as shown in FIG. 10.

As shown in FIG. 12 e, most of the remaining link inductor energy isthen transferred to this phase pair (into the motor), bringing the linkinductor current down to a low level. Switches S_(4u) and S_(6l) arethen turned off, causing the link inductor current again to be shuntedinto the link capacitor, raising the link reactance voltage to theslightly higher input line-to-line voltage on phases A_(i) and B_(i).Any excess link inductor energy is returned to the input. The linkinductor current then reverses, and the above described link reactancecurrent/voltage half-cycle repeats, but with switches that arecomplimentary to the first half-cycle, as is shown in FIGS. 12 f to 12j, and in Conduction Mode sequence 1300, and graphs 1301 and 1302. FIG.12 g shows the link reactance current exchange during the inductor'snegative current half-cycle, between conduction modes.

FIG. 11 summarizes the line and inductor current waveforms for a fewlink reactance cycles at and around the cycle of FIGS. 12 and 13.

Note that TWO power cycles occur during each link reactance cycle. InFIGS. 12 a-12 j, power is pumped IN during modes 1 and 2, extracted OUTduring modes 3 and 4, IN again during modes 5 and 6, and OUT againduring modes 7 and 8. The use of multi-leg drive produces eight modesrather than four, but even if polyphase input and/or output is not used,the presence of TWO successive in and out cycles during one cycle of theinductor current is notable.

As shown in FIGS. 12 a-12 j and FIG. 13, Conduction Mode sequence 1300,and in graphs 1301 and 1302, the link reactance continues to alternatebetween being connected to appropriate phase pairs and not connected atall, with current and power transfer occurring while connected, andvoltage ramping between phases while disconnected (as occurs between theclosely spaced dashed vertical lines of which 1303 in FIG. 13 is oneexample).

In general, when the controller 1500 deems it necessary, each switch isenabled, as is known in the art, by raising the voltage of the gate 204on switch 200 (shown in FIG. 2 a) above the corresponding terminal 205,as an example. Furthermore, each switch is enabled (in the preferred twogate version of the switch) while the portion of the switch that isbeing enabled is zero or reverse biased, such that the switch does notstart conduction until the changing link reactance voltage causes theswitch to become forward biased. Single gate AC switches may be used, aswith a one-way switch embedded in a four diode bridge rectifier, butachieving zero-voltage turn on is difficult, and conduction losses arehigher.

In FIG. 15, current through the inductor is sensed by sensor 1510, andthe FPGA 1500 integrates current flows to determine the current flowingin each phase (port) of the input and output portals. Phase voltagesensing circuits 1511 and 1512 allow the FPGA 1500 to control whichswitches to enable next, and when.

By contrast, note that the prior art structure of FIG. 8 has fourbi-directional switches on the input, and two on the output, with a linkinductor (no parallel capacitor) in between. That configuration is ahard switched buck-boost, and, like all prior buck-boost converters, ithas only 1 power transfer per link inductor cycle. Moreover, the linkinductor has a DC current component, unlike the converter of FIG. 1(which has NO average DC current, only AC).

FIG. 14 illustrates inductor current and voltage waveforms when theconverter of FIG. 1 is operating with reduced output voltage. Linkinductor 120 current from the input increases during modes 1 and 2 to amaximum level as for the full output voltage case of FIG. 13, but sincethe output voltage is half as high as for the full output voltage case,link inductor current decreases only half as quickly while dischargingto the output phases in modes 3 and 4. This will generally supply therequired output current before the link inductor current has fallen tozero or even near zero, such that there is a significant amount ofenergy left in the link inductor at the end of mode 4 in FIG. 14. Thisexcess energy is returned to the input in mode 5 and 1. Mode 1 in FIG.14 begins prior to the vertical axis. It can be seen that with zerooutput voltage, the current during modes 3 and 4 (and 7 and 8) will notdecrease at all, so that all link inductor energy is returned to theinput, allowing for the delivery of output current but with no powertransfer, as is required for current delivered at zero volts.

The Kim converter cannot return this excessive inductor energy back tothe input, as this requires bidirectional switches. Thus the Kimconverter must wait until the inductor energy drops to a sufficientlylow value, with the result that the link reactance frequency drops to avery low value as the output voltage approaches zero. This in turn cancause resonances with input and/or output filters. With zero voltageoutput, the Kim converter cannot function at all.

Note that the modes cited in Kim et al. differ somewhat from the modescited here. This is due to two reasons. The first is that, for brevity,the “capacitor ramping”, or “partial resonant” periods in this inventionare not all numbered, as there are eight of those periods. As indicatedin FIGS. 12 b and 12 g, voltage ramping periods preferably occur betweeneach successive pair of conduction modes. The second reason is that Kimet al. operate their converter such that it draws current from one inputphase pair per power cycle, and likewise delivers current to one phasepair per power cycle. This results in only two conduction modes per linkreactance cycle, since their converter only has one power cycle per linkreactance cycle. By contrast, FIGS. 12 a-12 j show current being drawnand delivered to both pairs of input and output phases, resulting in 4modes for each direction of link inductor current during a power cycle,for a total of 8 conduction modes since there are two power cycles perlink reactance cycle in the preferred embodiment. This distinction isnot dependent on the topology, as either three phase converter may beoperated in either 2 modes or 4 conduction modes per power cycle, butthe preferred method of operation is with 4 conduction modes per powercycle, as that minimizes input and output harmonics. For single phase ACor DC, it is preferred to have only two conduction modes per powercycle, or four modes per link reactance cycle, as there is only oneinput and output pair in that case. For mixed situations, as in theembodiment of FIG. 24 which converts between DC or single phase AC andthree phase AC, there may be 1 conduction mode for the DC interface, and2 for the three phase AC, for 3 conduction modes per power cycle, or 6modes per link reactance cycle. In any case, however, the two conductionmodes per power half-cycle for three phase operation together give asimilar power transfer effect as the singe conduction mode for singlephase AC or DC.

Control algorithms may use this ability of recycling inductor energy toadvantage in order to control current transfers, as is required by manyconverter control algorithms for vector or volts/Hz control. One suchpossible algorithm is explained in FIGS. 16 through 20. FIGS. 16, 17,and 20 show possible current profiles for the link inductor during apower cycle of positive current. This is for the case of only twoconduction modes per power cycle, as this invention uses for singlephase AC or DC. The power cycle for negative inductor current is themirror image of the cycles shown, as there are two power cycles perinductor cycle. Timing intervals T1, T2, T3, Tr1, and Tr2 are shown. T1is the time for the first conduction mode, when current is increasingfrom the input. T2 is the second conduction mode, in which the inductoris connected to the output, either decreasing in current for powertransfer to the output (positive power) as in FIGS. 16 and 17, orincreasing in current for power transfer from the output (negativepower) as in FIG. 20. T3 is the actually the first part of conductionmode 1 in which excess link inductor energy is either returned to theinput during positive power or delivered from output to input duringnegative power. Tr1 and Tr2 are the “partial resonant”, or “capacitorramping” times during which all switches are off and the voltage on thelink reactance is ramping. For three phase operation, intervals T1 andT2 are sub-divided, with T1 consisting of two conduction modes for thetwo input phase pairs from which current is drawn, and likewise for T2for delivery of current to the output phases. The relative times andinductor current levels determine the charge and therefore the relativecurrent levels among the phases. For three phase operation with zero ornear-zero power factor, T2 may subdivided into negative and positiveenergy transfer periods. Note that similar durations are used forramping the converter in BOTH directions. However, the ramping durationscan be different between input and output phases, as load draw variesdue to extrinsic circumstances. The charge time from the input can beheld constant, with the discharge time to the output varied to varyaverage output current (see FIG. 19). Excess link inductor energy(current) is returned to the input in T3. But all charge times andtransitions on the link reactance are perfectly symmetric about the zeropoints of voltage and current (see FIG. 13).

For the single phase AC and DC operation of FIGS. 16 through 20, theaverage output current is given by the equation at the bottom of FIGS.16, 17, and 20, with the “Charge over T2” given by the integral of thelink inductor current over the time interval of T2. For positive power,the peak link inductor current I1 may be held constant, while T2 isvaried to control average output current (I_(avg-out)). An algorithm tocalculate I_(avg-out) is shown in FIG. 18. For a given set of circuitparameters and input and output voltages, T2 (first column in FIG. 18)may be varied to control I_(avg-out) (sixth column). Resulting othertime intervals and power levels are also calculated. An input voltage of650 volts and an output voltage of 600 volts is used in FIG. 19. FIG. 19shows the results of the algorithm for other output voltages, with the650 volts input, as a function of T2, in micro-seconds (μsec). Anaverage (filtered) output current level of 26 amps is shown for the 650volt output curve with a T2 of 27 μsec, for a power output of 16.8 kW.Note that the link reactance frequency remains constant at 10 kHz forthe 650 volt output curve, regardless of T2 and I_(avg-out). For theother curves, with lower output voltage, frequency drops for loweroutput voltage, but never drops below 5 kHz even for zero output volts.Also note that I_(avg-out) for 0 volts goes to 55 amps for T2 of 50μsec, which is more than double I_(avg-out) at maximum power, eventhough maximum inductor current remains constant at 110 amps. For lowerconverter losses when lower output currents are commanded, thecontroller 1500 may be programmed to reduce T1, thereby reducing thepeak inductor current.

FIG. 19 also shows some specific drive parameters for the example 460VAC, 20 hp drive. The link inductor is 140 μH, and may be constructed asan air core copper wound inductor, with thin, flat, ribbon wire so as tohave a low ratio of AC to DC resistance from the skin effect, and woundlike a roll of tape. This configuration optimizes the inductance toresistance ratio of the inductor, and results in relatively highparasitic capacitance. Such a design cannot be used by hard switchedconverters, as this high parasitic capacitance causes high losses, butwith this invention the high parasitic capacitance is a benefit. Theramp, or parallel, link capacitance is comprised of two parallel AVX(FSV26B0104K—) 0.1 μF film capacitors capable of handling the RMScurrent load of about 25 amps. Peak inductor current is 110 amps.Commercially available reverse-blocking IGBT switches, IXYS part 40N12055 A, 1200 V, arranged in anti-parallel pairs may be used. In a standardhard switched application, such as a current source drive, this switchhas relatively high turn-on and reverse recovery losses caused by theslow reverse recovery time of the device, but when used in thisinvention, both turn-on and reverse recovery losses are negligible evenat a per device maximum switching frequency of 10 kHz and 110 amps peakcurrent. High RMS current capacitors from AVX (FFV3410406K), totaling 80μF line-to-neutral, may be used for the input and output capacitors. TheAltera Cyclone III FPGA may be used for the controller, implementing thealgorithms described above to control current flow, and using eithervector or Volts/Hz to control a 20 HP motor. Isolated power supplies,gate drivers, and digital isolators allow the FPGA to control the on-offstates of the IGBTs. Voltage and current sensing circuits, withanalog-digital interfaces to the FPGA, allow for precise switch timingto control current flow.

As may be surmised by those skilled in the art, the current resultingfrom the above described operation of the converter is, in manyapplications, controlled by controller 1500 to result in a sinusoidalvarying current from the input, normally in phase with the input voltageso as to produce a unity power factor on the input, and sinusoidallyvarying voltage and current on the motor, so as to operate the motor atthe highest possible efficiency and/or performance.

In those cases where the motor is acting as a generator, as may occurwhen the frequency applied to the motor via the converter is rapidlydecreased, the above described operating cycle is reversed, with currentbeing drawn from the motor phases and injected into the input phases.

In general, the input and output frequencies are substantially less thanthe frequency at which the link reactance is operated. For 60 Hz input,a typical operating frequency of the link reactance may be 10 kHz forlow voltage (230 to 690 VAC) drives and converters, and 1.5 kHz formedium voltage (2300 on up) drives and converters, with current pulsefrequencies twice those frequencies, or higher if multiple, synchronizedpower modules are used, as shown in FIG. 28. Input and Outputfrequencies may vary from zero (DC) to over well over 60 Hz, and mayeven be up to 20 kHz in audio amplifier applications.

Another embodiment of this invention is shown in FIG. 21, which shows asingle phase AC or DC to single phase AC to DC converter. Either or bothinput and output may be AC or DC, with no restrictions on the relativevoltages. If a portal is DC and may only have power flow either into orout of said portal, the switches applied to said portal may beuni-directional. An example of this is shown with the photovoltaic arrayof FIG. 23, which can only source power.

FIG. 22 shows another inventive embodiment, in a Flyback Converter. Herethe circuit of FIG. 21 has been modified, in that the link inductor isreplaced with a transformer 2200 that has a magnetizing inductance thatfunctions as the link inductor. Any embodiment of this invention may usesuch a transformer, which may be useful to provide full electricalisolation between portals, and/or to provide voltage and currenttranslation between portals, as is advantageous, for example, when afirst portal is a low voltage DC battery bank, and a second portal is120 volts AC, or when the converter is used as an active transformer.

In the embodiments of this invention shown in FIGS. 23 and 24, thenumber of portals attached to the link reactance is more than two,simply by using more switches to connect in additional portals to theinductor. As applied in the solar power system of FIG. 23, this allows asingle converter to direct power flow as needed between the portals,regardless of their polarity or magnitude. Thus, the solar photovoltaicarray may be at full power, 400 volts output, and delivering 50% of itspower to the battery bank at 320 volts, and the 50% to the house AC at230 VAC. Prior art requires at least two converters to handle thissituation, such as a DC-DC converter to transfer power from the solar PVarray to the batteries, and a separate DC-AC converter (inverter) totransfer power from the battery bank to the house, with consequentialhigher cost and electrical losses. The switches shown attached to thephotovoltaic power source need be only one-way since the source is DCand power can only flow out of the source, not in and out as with thebattery.

In the power converter of FIG. 24, as could be used for a hybridelectric vehicle, a first portal is the vehicle's battery bank, a secondportal is a variable voltage, variable speed generator run by thevehicle's engine, and a third portal is a motor for driving the wheelsof the vehicle. A fourth portal, not shown, could be external singlephase 230 VAC to charge the battery. Using this single converter, powermay be exchanged in any direction among the various portals. Forexample, the motor/generator may be at full output power, with 50% ofits power going to the battery, and 50% going to the wheel motor. Thenthe driver may depress the accelerator, at which time all of thegenerator power may be instantly applied to the wheel motor. Conversely,if the vehicle is braking, the full wheel motor power may be injectedinto the battery bank, with all of these modes using a single converter.

FIGS. 25 and 26 show half-bridge converter embodiments of this inventionfor single phase/DC and three phase AC applications, respectively. Thehalf-bridge embodiment requires only 50% as many switches, but resultsin 50% of the power transfer capability, and gives a ripple current inthe input and output filters which is about double that of the fullbridge implementations for a given power level.

FIG. 27 shows a sample embodiment as a single phase to three phasesynchronous motor drive, as may be used for driving a householdair-conditioner compressor at variable speed, with unity power factorand low harmonics input. Delivered power is pulsating at twice the inputpower frequency.

FIG. 28 shows a sample embodiment with dual, parallel power modules,with each module constructed as per the converter of FIG. 1, excludingthe I/O filtering. This arrangement may be advantageously used wheneverthe converter drive requirements exceed that obtainable from a singepower module and/or when redundancy is desired for reliability reasonsand/or to reduce I/O filter size, so as to reduce costs, losses, and toincrease available bandwidth. The power modules are best operated in amanner similar to multi-phase DC power supplies such that the linkreactance frequencies are identical and the current pulses drawn andsupplied to the input/output filters from each module are uniformlyspaced in time. This provides for a more uniform current draw andsupply, which may greatly reduce the per unit filtering requirement forthe converter. For example, going from one to two power modules,operated with a phase difference of 90 degrees referenced to each of themodules inductor/capacitor, produces a similar RMS current in the I/Ofilter capacitors, while doubling the ripple frequency on thosecapacitors. This allows the same I/O filter capacitors to be used, butfor twice the total power, so the per unit I/O filter capacitance isreduced by a factor 2. More importantly, since the ripple voltage isreduced by a factor of 2, and the frequency doubled, the input linereactance requirement is reduced by 4, allowing the total line reactormass to drop by 2, thereby reducing per unit line reactance requirementby a factor of 4.

FIG. 29 shows an embodiment as a three phase Power Line Conditioner, inwhich role it may act as an Active Filter and/or supply or absorbreactive power to control the power factor on the utility lines. If abattery, with series inductor to smooth current flow, is placed inparallel with the output capacitor 2901, the converter may then operateas an Uninterruptible Power Supply (UPS).

According to various disclosed embodiments, there is provided: aBuck-Boost Converter, comprising: an energy-transfer reactance; firstand second power portals, each with two or more ports by whichelectrical power is input from or output to said portals; first andsecond bridge switch arrays interposed between said reactance andrespective ones of said portals, and each comprising one bidirectionalswitching device for each said port of each said power portal; and alink switch, connected across said reactance in a crowbar configurationwhich includes resistance, to dissipate energy stored in said reactancewhen said link switch is on.

According to various disclosed embodiments, there is provided: aBuck-Boost Converter, comprising: an energy-transfer reactance; a firstbridge switch array comprising at least two bidirectional switchingdevices which are jointly connected to operatively connect at least oneterminal of said reactance to a power input, with reversible polarity ofconnection; a second bridge switch array comprising at least twobidirectional switching devices which are jointly connected tooperatively connect at least one terminal of said reactance to a poweroutput, with reversible polarity of connection; a link switch, connectedacross said reactance in a crowbar configuration which includesresistance, to dissipate energy stored in said reactance when said linkswitch is on; and wherein said first switch array drives said reactancewith a nonsinusoidal voltage waveform.

According to various disclosed embodiments, there is provided: aFull-Bridge Buck-Boost Converter, comprising: first and second fullbridge switch arrays, each comprising at least four bidirectionalswitching devices; a substantially parallel inductor-capacitorcombination symmetrically connected to be driven separately by eithersaid switch array; and a link switch, connected across saidinductor-capacitor combination in a crowbar configuration which includesresistance, to dissipate energy stored in said reactance when said linkswitch is on; one of said switch arrays being operatively connected to apower input, and the other thereof being operatively connected to supplya power output.

According to various disclosed embodiments, there is provided: aBuck-Boost Converter, comprising: first and second switch arrays, eachcomprising at least two bidirectional switching devices; a substantiallyparallel inductor-capacitor combination connected to each said switcharray; and a link switch, connected across said inductor-capacitorcombination in a crowbar configuration which includes resistance, todissipate energy stored in said reactance when said link switch is on;wherein a first one of said switch arrays is operatively connected to apower input, and is operated to drive power into said inductor-capacitorcombination with a non-sinusoidal waveform; and wherein a second one ofsaid switch arrays is operated to extract power from saidinductor-capacitor combination to an output.

According to various disclosed embodiments, there is provided: a powerconverter, comprising: an energy-transfer reactance comprising at leastone inductor, and operating at a primary AC magnetic field frequencywhich is less than half of the reactance's resonant frequency; a linkswitch, connected across said reactance in a crowbar configuration whichincludes resistance, to dissipate energy stored in said reactance whensaid link switch is on; an input switch array configured to drive ACcurrent through said reactance; and an output network switch arrayconnected to extract energy from said reactance; wherein said inputswitch array performs at least two drive operations, in the samedirection but from different sources, during a single half-cycle of saidreactance.

According to various disclosed embodiments, there is provided: a powerconverter, comprising: an energy-transfer reactance comprising at leastone inductor, and operating at a primary AC magnetic field frequencywhich is less than half of the reactance's resonant frequency; a linkswitch, connected across said inductor-capacitor combination in acrowbar configuration which includes resistance, to dissipate energystored in said reactance when said link switch is on; an input switcharray configured to drive current through said reactance; and an outputswitch array to extract energy from said reactance; wherein said inputswitch array performs at least two different drive operations atdifferent times during a single cycle of said reactance, and whereinsaid output switch array performs at least two different driveoperations at different times during a single cycle of said reactance.

According to various disclosed embodiments, there is provided: aBuck-Boost Converter, comprising: an energy-transfer reactancecomprising at least one inductor; and a link switch, connected acrosssaid inductor-capacitor combination in a crowbar configuration whichincludes resistance, to dissipate energy stored in said reactance whensaid link switch is on; a plurality of input switch arrays, each saidarray configured to drive AC current, with no average DC current,through said reactance; and a plurality of output switch arrays, eachconnected to extract energy from said reactance; said arrays having nomore than two switches driving or extracting energy from said reactanceat any given time; wherein said input switch arrays individually drivesaid reactance with a nonsinusoidal voltage waveform.

According to various disclosed embodiments, there is provided: a powerconversion circuit, comprising: an input stage which repeatedly, atvarious times, drives current into the parallel combination of aninductor and a capacitor, and immediately thereafter temporarilydisconnects said parallel combination from external connections, tothereby transfer some energy from said inductor to said capacitor;wherein said action of driving current is performed in opposite sensesand various times, and wherein said disconnecting operation is performedsubstantially identically for both directions of said step of drivingcurrent; a link switch, connected across said inductor-capacitorcombination in a crowbar configuration which includes resistance, todissipate energy stored in said reactance when said link switch is on;and an output stage which extracts energy from said parallelcombination, to thereby perform power conversion.

According to various disclosed embodiments, there is provided: a methodfor operating a Buck-Boost Converter, comprising the actions of: (a)operating a first bridge switch array, comprising bidirectionalswitching devices, to operatively connect at least one terminal of areactance to a power input, with polarity which reverses at differenttimes; (b) operating a second bridge switch array, comprisingbidirectional switching devices, to operatively connect at least oneterminal of said reactance to a power output, with polarity whichreverses at different times; and (c) under at least some overvoltageconditions, disconnecting said reactance from said power input or saidpower output or both, while also dumping energy from said reactancethrough a link switch which shunts said reactance; wherein said actions(a) and (b) are not performed simultaneously.

According to various disclosed embodiments, there is provided: a methodfor operating a Buck-Boost Converter, comprising the actions of:operating a first bridge switch array, comprising bidirectionalswitching devices, to operatively connect at least one terminal of asubstantially parallel inductor-capacitor combination to a power input,with polarity which reverses at different times; wherein said firstswitch array is operatively connected to a power input, and is operatedto drive power into said inductor-capacitor combination with anon-sinusoidal waveform; operating a second one of said switch arrays toextract power from said inductor-capacitor combination to an output; andunder at least some overvoltage conditions, disconnecting said reactancefrom said power input or said power output or both, while also dumpingenergy from said reactance through a link switch which shunts saidreactance.

According to various disclosed embodiments, there is provided: a methodfor operating a power converter, comprising the actions of: driving anenergy-transfer reactance with a full AC waveform, at a base frequencywhich is less than half the resonant frequency of said reactance;coupling power into said reactance, on each cycle thereof, with twodifferent drive phases, respectively supplied from two different legs ofa polyphase power source; and coupling power out of said reactance, oneach cycle thereof, with two different connection phases, respectivelydriving two different legs of a polyphase power output; and under atleast some overvoltage conditions, disconnecting said reactance fromsaid power input or said power output or both, while also dumping energyfrom said reactance through a link switch which shunts said reactance.

According to various disclosed embodiments, there is provided: a methodfor power conversion, comprising the actions of: driving anenergy-transfer reactance with a full AC waveform, at a base frequencywhich is less than half the resonant frequency of said reactance;coupling power into said reactance, on each cycle thereof, with twodifferent drive phases, respectively supplied from two different legs ofa polyphase power source; and extracting power from said reactance to anoutput; and under at least some overvoltage conditions, disconnectingsaid reactance from said power input or said power output or both, whilealso dumping energy from said reactance through a link switch whichshunts said reactance.

According to various disclosed embodiments, there is provided: aBuck-Boost power conversion method, comprising: operating an inputswitch array configured to drive AC current through an energy-transferreactance, at an average current magnitude which is more than 100 timesas great as the average DC current within said reactance; saidenergy-transfer reactance comprising at least one inductor; operating anoutput network to extract energy from said reactance; and under at leastsome overvoltage conditions, disconnecting said reactance from saidpower input or said power output or both, while also dumping energy fromsaid reactance through a link switch which shunts said reactance.

According to various disclosed embodiments, there is provided: a methodfor operating a power conversion circuit, comprising the steps ofrepeatedly, at various times: driving current into the parallelcombination of an inductor and a capacitor, and immediately thereaftertemporarily disconnecting said parallel combination from externalconnections, to thereby transfer some energy from said inductor to saidcapacitor; wherein said action of driving current is performed inopposite senses and various times, and wherein said disconnectingoperation is performed substantially identically for both directions ofsaid step of driving current; extracting energy from said parallelcombination, to thereby perform power conversion; and under at leastsome overvoltage conditions, disconnecting said reactance from saidpower input or said power output or both, while also dumping energy fromsaid reactance through a link switch which shunts said reactance.

According to various disclosed embodiments, there is provided: a methodfor operating a power conversion circuit, comprising the steps ofrepeatedly, at various times: a) driving current into the parallelcombination of an inductor and a capacitor, and immediately thereaftertemporarily disconnecting said parallel combination from externalconnections, to thereby transfer some energy from said inductor to saidcapacitor; b) coupling power out of said parallel combination, andimmediately thereafter temporarily disconnecting said parallelcombination from external connections, to thereby transfer some energyfrom said inductor to said capacitor; and (c) under at least someovervoltage conditions, disconnecting said reactance from said powerinput or said power output or both, while also dumping energy from saidreactance through a link switch which shunts said reactance; whereinsaid disconnecting operation, in said step a, is performed substantiallyidentically for both directions of said step of driving current; whereinsaid disconnecting operation, in said step b, is performed substantiallyidentically for both directions of said step of driving current.

According to various disclosed embodiments, there is provided: methods,circuits and systems for power conversion, using a universal multiportarchitecture. When a transient appears on the power input (which can be,for example, polyphase AC), the input and output switches are opened,and a crowbar switch shunts the inductance which is used for energytransfer. This prevents this inductance from creating an overvoltagewhen it is disconnected from outside lines.

MODIFICATIONS AND VARIATIONS

As will be recognized by those skilled in the art, the innovativeconcepts described in the present application can be modified and variedover a tremendous range of applications, and accordingly the scope ofpatented subject matter is not limited by any of the specific exemplaryteachings given. It is intended to embrace all such alternatives,modifications and variations that fall within the spirit and broad scopeof the appended claims.

In some preferred embodiments (but not necessarily in all), the linkreactance is driven with a nonsinusoidal waveform, unlike resonantconverters.

In some preferred embodiments (but not necessarily in all), capacitivereactances are used on both input and output sides.

In some preferred embodiments (but not necessarily in all), theswitching bridges are constructed with bidirectional semiconductordevices, and operated in a soft-switched mode.

In some preferred embodiments (but not necessarily in all), the inputswitching bridge is operated to provide two drive phases, from differentlegs of a polyphase input, during each cycle of the link reactance. Theoutput bridge is preferably operated analogously, to provide two outputconnection phases during each cycle of the reactance.

In some preferred embodiments (but not necessarily in all), the inputswitching bridge is operated to provide two drive phases, from differentlegs of a polyphase input, during each cycle of the link reactance. Theoutput bridge is preferably operated analogously, to provide two outputconnection phases during each cycle of the reactance.

In some preferred embodiments (but not necessarily in all), the linkreactance uses an inductor which is paralleled with a discretecapacitor, or which itself has a high parasitic capacitance.

None of the description in the present application should be read asimplying that any particular element, step, or function is an essentialelement which must be included in the claim scope: THE SCOPE OF PATENTEDSUBJECT MATTER IS DEFINED ONLY BY THE ALLOWED CLAIMS. Moreover, none ofthese claims are intended to invoke paragraph six of 35 USC section 112unless the exact words “means for” are followed by a participle.

The claims as filed are intended to be as comprehensive as possible, andNO subject matter is intentionally relinquished, dedicated, orabandoned.

1-32. (canceled)
 33. A power converter, comprising: an energy-transferreactance comprising at least one inductor, and operating at a primaryAC magnetic field frequency which is less than half of the reactance'sresonant frequency; a link switch, connected across said reactance in acrowbar configuration which includes resistance, to dissipate energystored in said reactance when said link switch is on; an input switcharray configured to drive AC current through said reactance; and anoutput network switch array connected to extract energy from saidreactance; wherein said input switch array performs at least two driveoperations, in the same direction but from different sources, during asingle half-cycle of said reactance.
 34. The converter of claim 33,wherein said switch arrays are full-bridge arrays.
 35. The converter ofclaim 33, wherein said reactance comprises a transformer.
 36. A powerconverter, comprising: an energy-transfer reactance comprising at leastone inductor, and operating at a primary AC magnetic field frequencywhich is less than half of the reactance's resonant frequency; a linkswitch, connected across said inductor-capacitor combination in acrowbar configuration which includes resistance, to dissipate energystored in said reactance when said link switch is on; an input switcharray configured to drive current through said reactance; and an outputswitch array to extract energy from said reactance; wherein said inputswitch array performs at least two different drive operations atdifferent times during a single cycle of said reactance, and whereinsaid output switch array performs at least two different driveoperations at different times during a single cycle of said reactance.37. The converter of claim 36, wherein said switch arrays arefull-bridge arrays.
 38. The converter of claim 36, wherein said firstarray connects said reactance to a power input which is shunted by acapacitor which provides a low-impedance voltage source thereat.
 39. Theconverter of claim 36, wherein said first array connects said reactanceto a power input which is shunted by a capacitor which provides alow-impedance voltage source thereat.
 40. The converter of claim 36,wherein said reactance comprises a transformer. 41.-60. (canceled)
 61. Amethod for operating a power converter, comprising the actions of:driving an energy-transfer reactance with a full AC waveform, at a basefrequency which is less than half the resonant frequency of saidreactance; coupling power into said reactance, on each cycle thereof,with two different drive phases, respectively supplied from twodifferent legs of a polyphase power source; and coupling power out ofsaid reactance, on each cycle thereof, with two different connectionphases, respectively driving two different legs of a polyphase poweroutput; and under at least some overvoltage conditions, disconnectingsaid reactance from said power input or said power output or both, whilealso dumping energy from said reactance through a link switch whichshunts said reactance.
 62. The method of claim 61, wherein said bridgearrays are symmetrically connected to said energy-transfer reactance.63. The method of claim 61, wherein said energy-transfer reactancecomprises a transformer.
 64. The method of claim 61, wherein saidenergy-transfer reactance comprises a parallel combination of aninductor with a capacitor.
 65. The method of claim 61, wherein saidreactance is driven at a base frequency which is less than half itsresonant frequency. 66-76. (canceled)